Field of the Invention
The present invention relates to telecommunication systems. More particularly, the present invention relates to transmission methods which make use of architectures of the type massive MIMO (Multiple-Input Multiple-Output) based on SC-FDE (Single-Carrier with Frequency Domain Equalization) schemes with large constellations that is compatible with low-cost, highly-efficient, nonlinear amplifiers, while allowing spatial multiplexing gains.
Description of Related Art
It is well-known that mm-wave (millimeter wave) communications have high potential for future wireless broadband systems. However, there are important challenges that need to be overcome when implementing those systems, both at the hardware level and due to the hostile propagation conditions. This is specially important for spectrally efficient communications based on large constellations, since the power requirements are higher, as well as the amplification difficulties, specially for large, dense constellations.
The use of mm-wave bands is expected to be a key feature of 5G (Fifth Generation) systems, since the large bands available allow significant capacity gains [1, 2, 3]. Moreover, the small wavelengths mean small antennas, together with small separation between antenna elements, allowing the implementation of massive MIMO (Multiple-Input and Multiple-Output) schemes. However, there are important challenges associated to the implementation of mm-wave systems, namely the high free space losses and difficult propagation conditions (e.g., small diffraction effects, huge losses due to obstacles) and difficulties at the hardware level. This combined with the large bit rates, means that the power requirements are much more severe than with conventional, sub-6 GHz, communications. This is particularly important for systems that also require high spectral efficiencies (i.e., large constellations), which not only have higher power requirements but are also associated to signals with higher envelope fluctuations and higher PAPR (Peak-to-Average Power Ratio), which leads to lower amplification efficiency [4]. Therefore, efficient power amplification is critical for mm-wave communications.
Another problem associated to broadband mm-wave communications is the frequency selectivity of the channel, since the good reflection properties of most materials at these frequencies lead to rich multipath environments. It is well-known that SC-FDE (Single-Carrier with Frequency Domain Equalization) is suitable for broadband communications over severely time-dispersive channels and severe power constraints [4, 5]. Although the PAPR problem is less serious with SC-FDE than with OFDM (Orthogonal Frequency-Division Multiplexing) schemes, a quasi-linear power amplifier is still required. It is known that class D or E amplifiers can have very high amplification efficiencies, together with low-cost implementations. However, since these amplifiers are strongly nonlinear, they should only be employed for signals with constant or quasi-constant envelope. OQPSK-type (Offset Quadrature Phase Shift Keying) signals can be decomposed as the sum of several linear OQPSK components [6], allowing good tradeoffs between reduced envelope fluctuations and a compact spectrum, including as special cases GMSK (Gaussian Minimum Shift Keying) and other CPM (Continuous Phase Modulation) schemes [7]).
The transmission method disclosed in the present application uses SC-FDE schemes, combined with offset modulations with large constellations, for high spectral efficiency mm-wave communications. To allow highly efficient, strongly nonlinear power amplifiers, the variable envelope signals associated to large constellations are decomposed as the sum of several polar components [8], each one modulated as a serial OQPSK signal [9] with reduced envelope fluctuations that is amplified and transmitted by a separate antenna within a massive MIMO scheme.
In the method disclosed in the present application it is employed a massive MIMO scheme with Nm×Nb antenna elements at the transmitter, arranged in Nm sets of Nb antenna elements or Nb sets of Nm antenna elements. As with conventional beamforming schemes Nb antenna elements are employed to define directive beams for spatial multiplexing purposes and/or interference management. However, the Nm elements associated to each of the Nb beamforming elements are employed to allow an efficient amplification of the signals associated to a large constellation, which is substantially different from conventional massive MIMO schemes.
In the method disclosed in the present application the data stream is split into Nu sub-streams that will be transmitted in parallel thanks to the spatial multiplexing capabilities of the antenna arrays employed for beamforming purposes (typically Nu<Nb). The data bits associated to each of the Nu sub-streams are mapped into a given constellation (e.g., a QAM (Quadrature Amplitude Modulation) constellation) characterized by the ordered set ={s0, s1, . . . , sM−1}, where M is the number of constellation symbols, following the rule(βn(μ−1),βn(μ−2), . . . ,βn(1),βn(0))sn∈,with (βn(μ−1),βn(μ−2), . . . ,βn(1),βn(0)) denoting the binary representation of n with μ=log2(M) bits. The constellations symbols are mapped in Nm polar components, that are the result of the decomposition of signal sn in M components given by
                              s          n                =                ⁢                              g            0                    +                                    g              1                        ⁢                          b              n                              (                0                )                                              +                                    g              2                        ⁢                          b              n                              (                1                )                                              +                                    g              3                        ⁢                          b              n                              (                0                )                                      ⁢                          b              n                              (                1                )                                              +                                    g              4                        ⁢                          b              n                              (                1                )                                              +                      (            …            )                                                            =                    ⁢                                                    ∑                                  i                  =                  0                                                  M                  -                  1                                            ⁢                                                          ⁢                                                g                  i                                ⁢                                                      ∏                                          m                      =                      0                                                              μ                      -                      1                                                        ⁢                                                                          ⁢                                                            (                                              b                        n                                                  (                          m                          )                                                                    )                                                              γ                                              m                        ,                        i                                                                                                                  =                                          ∑                                  i                  =                  0                                                  M                  -                  1                                            ⁢                                                          ⁢                                                g                  i                                ⁢                                  b                  n                                      eq                    ⁡                                          (                      i                      )                                                                                                          ,            
with (γμ−1,i γμ−2,i . . . γ1,i γ0,i) denoting the binary representation of i, bn(m)=(−1)βn(m) denoting the polar representation of the bit βn(m), bneq(i)=Πm=0μ−1(bn(m))γm,i denoting the ith polar component of sn and Nm is the number of non-zero gi coefficients of the referred decomposition equation.
Next each of the Nm polar components is modulated as a BPSK (Binary Phase Shift Keying) signal, being each of these Nm BPSK signals a serial representation of an OQPSK signal [9]. The corresponding signals can them be separately amplified by Nm nonlinear amplifiers before being transmitted by Nm×Nb antennas. Spatial multiplexing effects combined with beamforming gains are achieved since each of these Nm signals will be transmitted by Nb antenna elements, with appropriate phase shifts to provide directive beams.
In the method disclosed in the present application, combination losses are avoided, since the outputs of the Nm amplifiers are combined at the channel. Also the antenna vertical sub-array associated to the Nm antenna elements performs constellation shaping as seen by the receiver [10], and the Nb antenna elements placed horizontally, with spacing lower than or equal to λ/2 (where λ is the wavelength of the transmission frequency carrier), allow horizontal beams.
In [11] it is disclosed a method of transmitting data based on an M-QAM modulation with nonlinear amplification. The transmission technique proposed in this application follows a different approach because the transmitter employs Nm antennas in parallel, one for each of the BPSK signals of the serial format in which the multilevel constellation is decomposed. Besides that, each one of Nm BPSK signals is transmitted by a set of Nb antenna elements to provide directive beams.
In [12] are disclosed antenna arrays aimed to achieve a directive radiation pattern diagram, since the signals transmitted by the different antennas are correlated. The transmission structure proposed in this application, although based on two-dimensional antenna arrays, only the horizontal arrays achieve a directive radiation pattern since the signals transmitted by the Nm different vertical antennas are uncorrelated, contrarily to what happens in [12].
Although it uses a set of antennas, two kinds of directivity are introduced by the disclosed method in the present application. The first one is introduced by a constellation shaping as seen by the receiver achieved by the vertical arrangement of Nm antennas through a dependency on the configuration of constellations points on the desired direction of transmission. By contrast to the cases described in [12], the radiated power is not modified to maximize the radiated power in a given direction. The second one is introduced by the set of Nb antenna elements placed horizontally that assures a maximization of radiation pattern in the desired horizontal direction.
In the method disclosed in the present application, the signals are independent with uncorrelated bit streams in each set of Nm vertical antennas. In this case there is no spatial factor associated to the radiation pattern but only constellation shaping, contrarily to what was proposed in [12]. In the present application the various constituent signals suffer phase rotations according to their position on the set of Nm transmit antennas so that the constellation shape is only optimized in the desired direction.
The closest case consists on a transmission of Nm signals in parallel, similarly to what happens in a MIMO system, but unlike the MIMO wherein each signal is associated to a well-defined signal now each signal belongs to one of sub-constellations in which the constellation is decomposed. Also, unlike the MIMO without precoding, where at the receiver each signal can be received and decoded separately, the receiver for the uth data sub-stream in proposed method needs to combine all Nm received signals to generate the transmitted symbol and only after this operation may decode the transmitted bits.
Document [13] discloses a transmission method to increase the system's throughput. Contrarily to the method of the present disclosure there is no decomposition of the constellation into sub-constellations, neither constellation shaping together with horizontal beamforming. Moreover, the method of [13] uses a single antenna, since all signals after the multiplication by the spreading sequences are combined and transmitted by only one antenna.
In [14] there are disclosed methods for nonlinear encoding of 16-OQAM (Offset QAM) schemes, based on two nonlinear OQPSK signals specially designed to allow higher amplification efficiency due to its robustness against nonlinear distortions. The present transmission method applies to any constellation and do not have the format constrains associated to the pulse shapes and durations that affect the decomposition done in [14].
In [15] there are disclosed pragmatic FDE (Frequency Domain Equalization) receivers that have low complexity but allow excellent performance, even for large QAM constellations and highly non-uniform offset constellations. A more detailed study about the reason behind the poor performance of modulations equalized with conventional FDE schemes is also presented. The decomposition of the constellations of the present application is generic and is not restricted to the serial OQPSK format described in [14] and [15].
In [16] and [17] is disclosed a transmission technique where near constant-envelope multiuser precoding is employed with resort to a phase modulator, where independent data streams are directly modulated by a phase modulator. The method of the present disclosure must be not confused with the techniques of [16,17] neither with conventional beamforming where the signal emitted by each antenna has a large dynamic range. Thus, there is no decomposition of multilevel constellations into constant envelope sub-constellations, no beams are formed, and the signals emitted by each antenna are not formed by weighing of a symbol of a given constellation (e.g. M-QAM).
In [18] it is disclosed a transmission method that uses multiple and spatially separated antennas at both transmit and receive to offer spatial multiplexing and spatial diversity in the polarization domain by using a DP (Dual Polarized) antenna at both transmit and receive sides. DP is also applied in the transmission method described in [19] where orthogonal channels are assured through DP wave transmission, since vertically and horizontally polarized electromagnetic waves in many non-line-of-sight scenarios fade almost independently (with some degree of cross-coupling), and in line-of-sight scenarios the two transmitted orthogonal polarizations remain orthogonal through the channel. Contrarily to the transmission methods disclosed in [18], [19] and [20], the method disclosed in the present application do not needs dual polarization to achieve the orthogonality as well as the double directivity. None of the methods described in [18], [19] and [20] uses a decomposition of multilevel constellations into constant envelope sub-constellations, and the signals emitted by each antenna are not formed by weighing of a symbol of a given constellation (e.g. M-QAM). Furthermore, separation between users is achieved by two different kinds of directivities: one is the information directivity introduced by constellation shaping as seen by the receiver that depends on a specific angle of optimization and another is the directivity associated to the beamforming. Contrarily to the method described in [20], the separation between users is achieved not only by beamforming but also by changing the constellation shape according to an angle in which the constellation is optimized for each user. Therefore, even when the beams are close or interfere, separation between users is granted by the different constellation spatial arrangements assigned to the different users.